4047 Inverter

 

Power input block: U$1 connector
The raw DC supply comes in on this 2-pin connector. Pin P1 goes straight onto the main unregulated rail (feeding the transformer center tap, the regulator input, and the two bulk caps); pin P0 is the common ground return. As with the dimmer board, there's no series/reverse-protection diode here — correct polarity matters.

Bulk filtering: C2 + C3 (3300µF/35V each, in parallel)
Sitting across the raw rail (positive to the input node, negative to ground), these absorb the large pulsed currents the output MOSFETs pull from the supply every time they switch, keeping the center-tap rail from sagging or ringing hard at each transition.

Regulated supply block: IC2 (78XXS, populated as 7809) + R2 (220R)
IC2 drops the raw input down to a clean 9V. Rather than feeding IC1's VDD directly, the regulator's output goes through R2 (220R) first — a small series resistor into the 4047's supply pin. Notably, there's no local bypass/decoupling capacitor sitting right at IC1's VDD pin in this design (C2/C3 are the only caps on that side, and they're on the raw rail, not the regulated one) — worth keeping in mind since this rail sits next to a hard-switching power stage.

Oscillator block: IC1 (CD4047) + R1 (1k) + R4 (10k trimmer) + C1 (0.47µF)
C1 connects directly across the chip's C and R/C pins (the timing cap). The timing resistor is R1 in series with R4, run between the R and R/C pins — and R4's wiper is shorted back to one end, so like the dimmer board's trimmer, it's wired as a simple adjustable two-terminal resistor rather than a true potentiometer divider. R4 is the frequency-trim knob. The chip's AST/!AST/-T pins are tied together, and RES/RET/+T/VSS are all tied to ground — the standard hookup that puts the 4047 into continuous free-running astable mode (no external trigger). It free-runs and puts out two complementary square waves on Q and !Q.

Push-pull driver stage: Q1/Q3 + R5, and Q2/Q4 + R6
Each 4047 output feeds a discrete complementary buffer rather than driving the power MOSFETs directly. Q drives Q1 (NPN, SS8050) and Q3 (PNP, SS8550) through the shared 10R base resistor R5; !Q drives Q2/Q4 the same way through R6. In each pair, the NPN's collector sits on the raw +rail and the PNP's collector sits on ground, with both emitters tied together as the output node — a classic discrete totem-pole buffer, needed because the 4047's own outputs can't source/sink enough current to slew the power MOSFETs' gate capacitance quickly.

Gate-drive block: R7–R10 (10R each) + R11/R12 (10k each)
Each power MOSFET gets its own series gate resistor (R7→Q5, R8→Q6, R9→Q7, R10→Q8) even though the MOSFETs are paralleled in pairs — correct practice, since it damps ringing and stops parasitic oscillation between paralleled devices sharing one driver node. R11 and R12 are shared 10k pulldowns on each pair's gate node, holding the MOSFETs off if the driver stage output is ever floating.

Power switching / output stage: Q5+Q6 and Q7+Q8 (IRF3205) + TERM1/TERM2/TERM-CT
Q5 and Q6 are paralleled (drains tied together, sources tied to ground) and their common drain feeds TERM2. Q7 and Q8 are paralleled the same way, feeding TERM1. TERM-CT ties directly to the raw +rail. Read together, this is a push-pull drive into a center-tapped transformer primary: TERM-CT is the center tap, TERM1/TERM2 are the two winding ends, and each end gets alternately pulled to ground by its MOSFET pair while the center tap stays at the supply rail — one classic way to turn a DC rail into an AC square wave. Small note: Q5–Q8 are built on the "IRF740" library deviceset with the value field overridden to "IRF3205" — worth double-checking that the footprint/ratings actually match the part you intend to populate, since those are quite different devices electrically.

Indicator block: R3 (10k) + LED1
R3 feeds LED1 from the regulated 9V rail to ground — a simple power-on indicator, separate from R2 (which only feeds the oscillator's VDD).

How it comes together
IC1 free-runs at a rate set by R1+R4 and C1, producing two antiphase square waves. Each phase is buffered through a discrete push-pull driver (Q1/Q3 or Q2/Q4) to get enough peak current to switch a pair of paralleled IRF3205s quickly. Those MOSFET pairs alternately pull each end of the transformer's center-tapped primary to ground while the center tap sits on the raw DC rail — driving the transformer push-pull at the oscillator's frequency. Given the topology (center-tapped push-pull into a transformer, big paralleled MOSFETs, large bulk caps), this reads as a DC-to-AC square-wave inverter, and R4's trim range is almost certainly meant to dial the switching frequency in to match a target mains frequency (50/60Hz) rather than anything higher — worth confirming against the transformer you're actually driving, since running it off-frequency from what the transformer/load expects is the main thing to get right here.

you-tube video with all files :
https://www.youtube.com/watch?v=C7B8RIV5g_g
cd4047 IRF3205 12V 500W 60Hz Inverter

113 kHz Piezoelectric Disc Driver

Power input block: 15V connector
15V comes in directly onto the main rail and ground, with no series protection diode. Correct-polarity connection matters here, since there's nothing to catch a reversed connection.

Regulated supply block: IC2 (78XX-series) + C4, C5, C6, C7
IC2 steps the 15V rail down to a regulated output that feeds the 555's VCC and reset pins, keeping the timer's supply separate from the higher-current switching rail. C7 (0.1µF) and C4 (100µF) bypass/bulk the regulator's input; C6 (0.1µF) and C5 (100µF) do the same on the output side. Note: the schematic uses a generic 78XX symbol with no specific voltage value set, so the exact regulated voltage isn't confirmable from the file alone — worth checking against your actual populated part.

Oscillator block: IC1 (NE555) + R4 + C3 + C2
This is a self-triggering relaxation oscillator: threshold and trigger are tied together, and that node charges/discharges through the trimmer R4 (10k) directly from the 555's own output, with C3 (10nF) as the timing cap. Discharge is unused. C2 (0.1µF) is the standard control-voltage bypass. R4 is your frequency-trim knob, meant to bring the 555's free-running rate into alignment with the LC tank's resonant frequency below.

Gate-drive block: R1 + R3
R1 (10R) is the gate series resistor from the 555's output, limiting instantaneous gate-charge current and damping ringing in the gate-drive loop. R3 (10k) pulls the gate to ground, keeping Q1 reliably off if the gate is ever left floating.

Resonant output block: Q1 + L1 (330µH) + C1 (0.1µF)
Q1's source is grounded, and its drain sits on a node shared by L1 and C1. L1's other end is wired directly to the 15V rail. When Q1 turns on, current builds through L1 from the rail; when Q1 turns off, that current can't stop instantly, so the drain voltage swings upward and the LC network (L1 together with C1 and the piezo's capacitance, which sit effectively in series from the drain's perspective) goes into its resonant transition — an approximately sinusoidal ring, with the exact shape depending on switching timing, inductor Q, piezo impedance, and how closely the 555's frequency tracks the loaded resonance. C1 couples that ringing drain node out to the PIZO connector.

Indicator block: R2 + LED1
R2 (10k) feeds LED1 from the regulated rail to ground — a simple "board is powered" indicator, unrelated to the oscillator or output stage.

How it comes together
The 555 free-runs at a rate set by R4 and C3, tuned close to the LC tank's resonance. Each cycle, Q1 switches, energy transfers into the tank, the tank rings near its natural frequency, and the direct connection to the 15V rail replenishes the energy the piezo dissipates as it vibrates. It's open-loop throughout — nothing about the piezo's or drain's actual waveform feeds back into the 555; the design's job is simply getting the oscillator frequency and the loaded resonance close enough together that the switching drives the tank efficiently. Since L1 is now 330µH (a real change from whatever value was there when this was last tuned), the resonant frequency has shifted, and the 555's rate via R4/C3 should be re-checked against the tank's new resonance rather than assumed to still be correct.
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you tube video link with all files :
https://www.youtube.com/watch?v=kNJp2BRtzR4
113Khz Water Atomizer Circuit + PCB

Temperature-Controlled Fan

 Temperature-Controlled Fan

Here's the walkthrough with those two refinements folded in:

Power block: 12V-IN, +12V, GND
Power comes in through the 12V-IN connector onto the +12V and GND rails. Everything below is referenced to these two rails.

Reference block: R1 — the threshold-set divider
R1 is a 10k trimmer wired end-to-end between +12V and GND, with its wiper feeding the op-amp's non-inverting input. This just creates an adjustable reference voltage — turning the trimmer sets exactly what voltage the comparator will treat as "the trip point." Nothing here reacts to temperature; it's a fixed (but adjustable) reference.

Sensing block: R2 + R6 — the temperature divider
R2 (4.7k) sits between +12V and the sensing junction; R6 (the NTC thermistor) sits between that same junction and GND. That junction feeds the op-amp's inverting input. As temperature rises, R6's resistance drops, so more of the +12V drop falls across R2 — meaning the junction voltage (the inverting input) falls as it gets hotter.

Comparator block: IC1 (LM358)
The op-amp is wired open-loop, so it acts as a comparator: reference voltage on the non-inverting input, temperature-dependent voltage on the inverting input. As temperature climbs and the inverting input drops below the reference, the output swings high. Below threshold, output stays low. There's no feedback path back into either input here, which is worth noting — this comparator has no built-in hysteresis, so right around the trip point it could in principle chatter a little rather than switch cleanly.

Zener threshold/offset block: D2 + R4
The comparator's output doesn't drive the transistor directly — it goes through a 3V zener (D2) first, then R4 (1k) into the base. The zener's cathode faces the op-amp output; its anode faces R4. When the output is low, the zener simply blocks — no base current, transistor stays off. When the output swings high enough to exceed the zener's breakdown voltage, it conducts and passes the excess through R4 into the base. This builds in a fixed voltage threshold before the transistor gets any drive, rather than reacting to the very first millivolt of comparator swing — though it's worth keeping in mind that a zener's breakdown voltage isn't a precise, fixed number; it shifts a bit with current, temperature, and part tolerance, so "3V" here is approximate rather than exact.

Switching block: Q1 (BC547)
Q1's emitter is grounded, base comes from the zener stage above, and its collector is the relay coil's low side. When base current flows, Q1 turns on and pulls the coil's low side down toward ground, completing the circuit (the coil's other end is tied straight to +12V) and energizing the relay.

Relay block: K1 + flyback diode D1
D1 (1N4004) sits across the coil — cathode on +12V, anode on the switched (collector) side — as a flyback diode, clamping the inductive kick when Q1 turns off so it doesn't damage the transistor. On the switch side, the relay's common pole is tied to +12V, and its normally-open contact feeds the fan. So energizing the coil is what actually connects +12V through to the fan.

Output block: FAN connector
The fan's positive side comes from the relay contact (so it only gets power when the relay is energized); its return goes straight to GND. So: hot enough → comparator trips → transistor pulls the relay coil low → relay closes → fan gets +12V and spins.

Indicator: LED1 + R3
This is on its own separate branch — R3 (1k) from +12V straight into the LED, cathode to GND. It has no connection to the comparator, transistor, or relay at all, so it's simply a "power is on" indicator, not a "fan is running" indicator.

Net effect
Temperature rises above setpoint: NTC resistance falls → sensing voltage falls → inverting input drops below the reference → output goes high → Q1 turns on → relay energizes → fan gets +12V.
Temperature falls back below setpoint: NTC resistance rises → sensing voltage rises → inverting input goes back above the reference → output goes low → Q1 turns off → relay de-energizes → fan stops.

Unlike your latch circuit, this one has no memory — the fan just follows the thermistor directly, with no feedback path from the output back into the sensing or reference network. That also means no deliberate hysteresis: if the temperature sits right at the threshold, the relay can potentially chatter.
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old you tube link with files : 
https://www.youtube.com/watch?v=ngJHaid4X3Q
temperature activated fan (උෂ්ණත්ව සංවේදී විදුලි පංකාව)

 

Understanding a Thermistor-Based Temperature Alarm Latch

 Understanding a Thermistor-Based Temperature Alarm Latch

 

 

Power block: PWR, +5V, GND
Power comes in through the PWR JST connector onto the +5V and GND rails. Every other block below is referenced to these two rails.

Sensing block: R19 + R21 — the resistor divider
R19 (the NTC thermistor) sits between +5V and the divider junction. R21 (the trimmer pot, wired as a two-terminal rheostat) sits between that same junction and GND. So the divider is: +5V → R19 → divider junction → R21 → GND. As temperature rises, R19's resistance drops, so the divider junction's voltage rises. R21 sets how far it needs to rise before the next stage reacts — that's your trip-point adjustment. D2's anode taps directly into this divider junction.

Trigger block: D2
D2's cathode sits on the latch's output node — the same node shared by Q2's collector, R10, R16, and C3. Normally that node sits low (latch off), so as the divider junction climbs with temperature, once it gets about a diode-drop above the output node, D2 starts conducting and pushes a small current into it. This is the "detect" event — a small trickle of current appearing there the moment the voltage condition is met. D2 is really acting as the voltage-sensitive trigger interface; the transistors that follow provide the actual amplification and memory.

What that trigger current does — through R10 into T2's base
That injected current at the output node has one meaningful path onward: through R10 (4.7K) into T2's base, which it shares with reset switch S2. So the small trigger current nudges T2 slightly into conduction.

Latch block: T2 + Q2 — the regenerative pair
This is the core mechanism:

• T2 (NPN) starts turning on → pulls its collector down.
• T2's collector feeds Q2's base through R15 (2.2K), so as it drops, Q2's base gets pulled down → Q2 (PNP) turns on.
• Q2's emitter is on +5V, so turning on pulls its collector — which is that same output node D2 feeds into — up toward +5V.
• That rise pushes more current through R10 into T2's base than the thermistor alone ever could.

That's the loop: T2 on → Q2 on → output node rises → drives T2 harder. This is regenerative switching — exactly how fast it snaps over depends on the transistor characteristics, capacitances, and resistor values, but the direction is unambiguous: once it starts, it runs away to fully on. Once latched, D2 and the thermistor stop mattering — the output node sits higher than the divider junction, so D2 goes reverse-biased and is out of the picture. The circuit is now holding the trip state purely through this feedback, not through any capacitor.

Supporting resistors and caps on the latch:

• R17 (2.2K) is the pull-up from +5V to T2's collector. With T2 off it holds T2's collector, and therefore Q2's base, high — keeping Q2 off. When T2 turns on, it pulls that node low, pulling Q2's base low and turning Q2 on.
• C4 (33µF, +5V to T2's collector) and C3 (1µF, output node to GND) provide filtering and transient stabilization around the regenerative nodes, making the circuit less sensitive to rapid voltage fluctuations.
• R16 (1K, output node to GND) gives that node a defined path to ground when Q2 is off, and also sets how much current Q2 has to supply once it turns on.

Output block: SG2 + R18, and LED2 + R13/R14
Once latched, the output node sits near +5V. From there:

• R18 (10R) current-limits into the buzzer SG2, whose return goes to GND — this sounds the alarm.
• Separately, T2's emitter node drives the indicator: R13 (330R) feeds LED2's anode, LED2's cathode goes to GND, lighting it up. R14 (10K) sits from that same emitter node to GND in parallel, giving it a defined load/pull-down path too.

Reset block: S2
S2 connects T2's base directly to GND when pressed, forcing T2 off regardless of the latch state. With T2 off, R17 pulls its collector back up, releasing the pull-down on Q2's base, so Q2 turns off too, and the output node falls back low — buzzer and LED both go out. The important caveat: resetting doesn't remove the temperature condition. If the divider junction is still sufficiently above the output node, D2 conducts again the moment you release S2, and the regenerative process starts right back up.
-
old youtube video which contain the demo+circuit+pcb files :
https://www.youtube.com/watch?v=_q9TxOG8NQc
fire detector circuit (උෂ්ණත්ව සංවේදී පරිපථය).

practical diode-protection cheat sheet

 

1. Polarity Protection

Topology	How it works	Trade-off
Series diode	Diode in line with supply; wrong polarity → blocked	Simple, but costs a forward-voltage drop (less with Schottky)
Parallel crowbar diode	Diode across supply, normally reverse-biased; wrong polarity → conducts hard and blows a fuse	Robust, zero drop in normal operation, but needs a fuse/current limiter sized correctly
MOSFET "ideal diode"	MOSFET replaces the series diode, driven by a small controller or gate-charge circuit	Near-zero voltage drop; standard in battery and automotive gear
Bridge rectifier	Full bridge in the power path; input polarity no longer matters	~2 diode drops in the current path, but works for AC or DC-either-way inputs

2. Overvoltage Protection

Topology	How it works	Trade-off
Zener clamp	Zener from signal to ground; conducts once voltage exceeds Vz	Simple, but limited power handling and soft knee
TVS diode	Avalanche diode built specifically for transient suppression (ESD, inductive spikes, automotive)	Fast response, higher surge capability than a Zener; unidirectional or bidirectional
Avalanche diode	Diode/Zener/TVS chosen to break down at a controlled voltage	Common for clamping inductive spikes from motors, solenoids, relay coils
Steering-diode clamp	Diodes route overvoltage to a nearby rail (VCC or ground) instead of dissipating it locally	Good for ADC/analog input protection, usually paired with a series resistor

3. Negative-Voltage / Bidirectional Protection

Topology	How it works	Trade-off
Two-diode input clamp	One diode clamps to VCC, one clamps to ground — most MCU pins have this built in internally	Cheap, but only protects within a diode drop above/below the rails
Back-to-back diodes	Two diodes oriented oppositely across a signal line, limiting both polarities symmetrically	Good for bidirectional signal protection against small over/under swings
Bidirectional TVS	Single TVS package that clamps both polarities	Best combined ESD + transient protection for signal lines

4. Inductive Load / Switching Spike Protection

Topology	How it works	Trade-off
Flyback diode	Diode across the coil (relay, solenoid, motor) gives stored energy a path when the switch opens	Protects the driving transistor/MOSFET; slows relay release time
Zener + flyback diode	Zener in series with (or replacing) the flyback diode lets the coil collapse at a higher voltage	Faster release/turn-off, useful when timing matters
RC snubber	Resistor + capacitor across the switching element	Reduces spike amplitude and EMI, doesn't fully clamp
RCD clamp	Resistor + capacitor + diode network	Common in SMPS to absorb transformer leakage-inductance energy

5. Multiple Power Sources

Topology	How it works	Trade-off
Diode OR-ing	Each supply feeds the load through its own diode; only the highest-voltage source conducts	Simple, but a full diode drop lost from each source
Ideal-diode MOSFET OR-ing	MOSFET version of the above	Much lower loss, more complex control circuitry
Battery isolation diode	Same principle applied specifically to prevent one battery from charging/discharging another	Same trade-offs as diode OR-ing

6. Dedicated Protection ICs

Topology	How it works	Trade-off
ESD diode array	IC containing multiple low-capacitance diodes tied to a common rail structure	Used for USB, CAN, HDMI, RS-485, GPIO — designed to shunt ESD without loading the signal
Diode matrix / steering network	Multiple diodes route abnormal voltages from several inputs toward safe rails	Used in keyboards, multiplexed inputs, interface protection

7. Two-Layer / Combined Schemes

Topology	How it works
Series diode + Zener	Series diode handles reverse polarity, Zener handles overvoltage — simple two-stage protection for low-power circuits
Diode + fuse crowbar	Crowbar diode conducts on reverse polarity, fuse opens before damage — prioritizes simplicity over efficiency

The key distinction

Ordinary diodes, Zeners, TVS diodes, and MOSFET "ideal diodes" solve different failure mechanisms and aren't interchangeable:

• Ordinary diode → one-way current blocking (polarity, OR-ing)
• Zener → moderate-power voltage limiting at a fixed level
• TVS → fast, high-surge transient absorption
• MOSFET ideal diode → polarity/OR-ing protection with near-zero loss

Picking the wrong one for the job (e.g., a Zener where you need TVS-level surge handling, or a series diode where you need MOSFET-level efficiency) is the most common design mistake in this space.

PROTECTION CIRCUIT LIMITATIONS

Protection circuits are not magic shields. Every protection method has a voltage, current, energy, time, and frequency limit.

Series diode

Protects against reverse polarity, but does not protect against overvoltage.

The diode itself must also be rated for the maximum forward current and reverse voltage.

Crowbar diode + fuse

Protects against reverse-polarity connection, but depends on the fuse opening quickly enough.

The diode must survive the fault current until the fuse operates. A wrong fuse rating can make the protection ineffective or damage the diode.

MOSFET ideal diode

Provides very low-loss reverse-polarity or power OR-ing protection, but the MOSFET still has limits for voltage, current, power dissipation and transient energy.

It is not automatically an overvoltage protector.

Bridge rectifier

Makes input polarity irrelevant, but introduces approximately two diode forward-voltage drops in the current path.

It also does not inherently protect against overvoltage or current overload.

Zener diode

A Zener is mainly a voltage clamp, not a universal surge protector.

Its ability to absorb a fault depends on its power rating and the duration of the event. A series resistor or current-limiting element is often essential.

TVS diode

A TVS is designed to absorb short-duration transients, not unlimited continuous overvoltage.

If an excessive voltage remains continuously, the TVS can overheat and fail. The TVS must therefore be selected according to standoff voltage, clamping voltage, peak pulse current and pulse energy.

Steering-diode clamp

Steering diodes can safely redirect an abnormal signal only if the destination rail can absorb the resulting current.

Without current limiting, such as a suitable series resistor, the protection diode or the supply rail itself can be damaged.

Flyback diode

Protects a switching transistor from the voltage generated by an inductive load when it is switched OFF.

However, a conventional flyback diode also makes the coil current decay more slowly. It therefore may not be suitable when fast release is required.

Zener + flyback

Allows a higher voltage across the coil during turn-off, producing faster energy decay.

But the Zener must be rated for the resulting pulse current and energy. The switching transistor must also be rated for the higher voltage.

RC snubber

Reduces switching spikes and EMI, but does not guarantee that the voltage will remain below a particular safe level.

Its effectiveness depends strongly on the resistor, capacitor, switching frequency and parasitic inductance of the actual circuit.

RCD clamp

Useful for absorbing controlled amounts of leakage-inductance energy, especially in switching power supplies.

The components must be designed for the actual switching voltage, current, frequency and energy. An incorrectly designed RCD clamp can itself become a source of heating or failure.

Diode OR-ing

Prevents one power source from directly feeding another, but the diode introduces power loss.

It also does not automatically provide current limiting, short-circuit protection or overvoltage protection.

ESD diode array

Designed mainly for short, fast ESD/transient events.

It is not intended to continuously dissipate a large overvoltage. Signal-line capacitance and leakage current must also be considered, especially with high-speed interfaces.

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THE MOST IMPORTANT RULE

When selecting a protection component, ask five questions:

1. What is the abnormal condition?
Reverse polarity? Overvoltage? ESD? Inductive spike? Short circuit?

2. How much voltage can appear?
The protection device must survive the actual voltage, not just the nominal operating voltage.

3. How much current can flow?
A clamp without current limiting can become the thing that fails.

4. How much energy must be absorbed?
A 1 kV pulse lasting a few nanoseconds and a 30 V fault lasting several seconds are completely different protection problems.

5. What happens if the protection device itself fails?
A TVS, Zener or crowbar diode may fail short or open. The surrounding fuse, resistor, MOSFET or power supply should be considered as part of the complete protection system.

In one sentence:

Protection design is not simply about choosing a diode with the right voltage rating — it is about controlling voltage, current, energy and time during a specific failure event.